Pop. Pop. Pop. Pop. It's not the Fourth of July. It could be your own dc-dc design blowing up, worries our power supply expert, Sanjaya Maniktala. You do want to check out your design under a variety of adverse conditions, he suggests. Fault management is the key to designing power supplies that last.
Now that you finally got your dc-dc converter working, is it really time to start popping the champagne? Shouldn't you at least wait to see whether it even survives the very first abnormal condition that comes its way? You don't really want to be the one to have to decipher (in your rather understandably hazy and exulted condition) whether that second pop came from the adjoining bottle or from your very own power supply. Fault management is the key to designing power supplies that last. Unfortunately, all this also has the immediately sobering potential to completely unmask the full extent of your design capabilities! Therefore, ensure that your current limiting is up to the task before you begin to celebrate.
In a typical high-power off-line supply, fault management requires a clear understanding of the required shutdown and subsequent recovery sequencing of the various constituent stages: the power factor correction stage, the auxiliary (house-keeping) power supply and the power trains. This is complex for sure. Therefore, in single-chip converters, we tend to think that the task of surviving abnormal conditions is trivial. But not so fast buster! A great deal depends on how quickly the current limiting acts. Also, whether it is set correctly to start with. And whether it can even do its expected job. (For the latter aspect we will delve into the subtleties of 'frequency foldback' in our next column. Here we first do all the necessary spadework.)
In the high-voltage off-line power industry, the underlying design philosophy is to size the transformer as per the current limit. Which means that the transformer should not saturate even if the current hits the limit, as it usually will during any normal power-up or under shorts or overloads. Of course it is always good to have the flexibility to set the current limit 'correctly' - to be just able to meet the required holdup time, step load capability, or any specified transient peak load requirement (as for disk drive motors, incandescent lamp loads etc). So though the thickness of the copper windings is certainly determined only by the continuous operating current (long-term heating effects), the actual physical size of the magnetic core must be set strictly according to the current limit alone. And this may or may not be related to the continuous operating current, as could easily happen if we use some popular integrated switcher families which provide only a discrete range of built-in fixed current limits.
Implicit in this design strategy is the realistic realization that we just cannot set a given current limit, and expect it to be enforced fast enough to be able to save the switch if the transformer starts saturating, even momentarily. While this was blatantly obvious for bipolar transistors in the past, though the situation has improved with the advent of FETs, it is not enough to have changed this basic design philosophy. At least not for high-voltage applications.
The design philosophy prevalent within the low-voltage dc-dc semiconductor industry is so vastly different from that of the off-line power industry, that transmigratory power engineers (nomads like me in an eternal search for a meaningful home) have a hard time reconciling to initially. Here too, we can have families of integrated switchers with fixed current limits, but we almost invariably end up totally ignoring the current limit (just so long as it is high enough to guarantee the desired output power). So we size the inductor according to the continuous operating load current --- no more, no less. For example we may use a switcher with a 5A current limit for a 2A application, and use only an inductor rated for 2A. We could also use the same switcher for a 4A application and we would then use a 4A inductor. Here we are assuming that since the switch is obviously OK with a current of 5A, then even if we use a 2A inductor, and it saturates momentarily, the current limit circuitry is capable of acting fast enough to immediately turn the transistor OFF the exact moment that it attempts to exceed 5A. So the switch can never be destroyed, since its legal limit is assiduously enforceable.
Now this scenario is clearly more likely to be true for FETs than bipolars, because the former have virtually no storage/delay times and can react almost immediately. But "almost" is not always good enough. Not in power. For example, if the input voltage is raised, the saturating inductor's current may ramp up so steeply, that even during the minimum ON time (minimum pulse width), the current may spike up high enough to damage the switch. In this case, we should ensure that the inductor is not saturating at the moment the current attempts to exceed the set current limit. That gives the circuitry enough time to safely turn the switch OFF.
While testing and evaluating the LM2590HV to LM2593HV family of high-voltage devices we observed some failures if we used very small sized inductors when the input voltage was higher than 40V. Thus our official recommendation in the datasheets for the inductor selection is somewhat of a cross between the two design philosophies talked about above - we have stated that if the input is less than 40V, the inductor should be sized as per the continuous rated load current, but if the input exceeds 40V, we should size the inductor as per the fixed current limit of the device (disregarding the load current totally).
We should also keep in mind that the above mentioned devices are bipolar-based integrated switchers. We can expect the situation to be better if we use integrated devices which use FET switches. On the other hand, if we are using controllers, then even with FET switches we should be cautious, because the switching speeds and corresponding delays are likely to be worse than with integrated switchers. It all boils down to our ability to set the current limit accurately and to enforce it fast enough. It is not a trivial proposition.
The reader may wonder how we ensure rock-solid reliability for low-voltage applications if we disregard the current limit altogether, as we effectively seem to be doing above. Well in reality we don't, but this aspect is cleverly hidden from most users, though not deliberately so, and certainly not to their disadvantage! Most integrated switcher families are actually virtually 'bullet-proof'. And one of the key methods to ensure this is by providing a second level of current limit protection,> usually not even mentioned in the datasheet, since it is rarely encountered and almost completely transparent to the user. But this second limit is typically about 20 to 30 percent higher than the first (declared) current limit. If this is ever encountered, as with severely saturating inductors and high input voltages, the IC is designed to enter a foldback condition. As we said, more on this subtlety the next time!!
Till then don't forget to drop me a line at firstname.lastname@example.org and do copy Steve at email@example.com.