SLIC Supplies

Subscriber-line interface cards (SLICs) provide the interface between the telephone service provider and the telephone handset in your home. They operate in two main modes: on-hook refers to when the handset is idle and waiting for a signal that indicates someone wants to make a connection, and off-hook refers to when the handset is active and the user is trying to complete a connection. Telephone-system voltages are traditionally negative to prevent electromigration from eroding the installed copper wiring.

Telephone systems require certain special voltages that vary from application to application and country to country. This article presents circuits for deriving these voltages from commonly available supply voltages. Table 1 summarizes the input and output characteristics of the circuits.

The on-hook voltage, which generates the ringer voltage, is typically -72 V in the United States and as high as -150 V in other countries. The ringer voltage, a 20- to 60-Hz sinusoid, drives an electromechanical bell in the handset that can be located far from the central office (CO). The off-hook voltage is typically -48 V in the United States, although some localized systems use -24 V. This voltage powers the system during voice communications. The CO provides power for the telephone system independently from the electric utility, allowing the telephone to work during a power outage.

With advances in data-communication technology, companies are incorporating voice service with data service to provide integrated communication systems. Such systems require SLIC functions to maintain compatibility with legacy equipment. The following circuits demonstrate techniques for generating SLIC voltages from commonly available voltages. All of the circuits are based on a transformer flyback topology using a MAX668 boost controller. This topology achieves compact magnetics and flexible output voltages.

-48-V Output from +5-V Supply

The circuit in Figure 1 generates -48 V at 300 mA for customer premises equipment (CPE) or client-side equipment from a +4.5-V minimum input. The input voltage also is the gate-drive voltage for the MAX668 (U1), limiting the input voltage to +5.5 V (max). The MOSFET switch (Q1) presents more gate capacitance than the controller can drive efficiently; therefore, complementary emitter followers (Q2, Q3) buffer the gate-drive output.

Q1 is selected to switch 9-A peak current with VGS = 3.8 V (4.5 VIN-VBE). A snubber for the flyback voltage is not necessary because the breakdown voltage of Q1 is nine times the voltage reflected back from the secondary to primary transformer winding. R7 and C7 filter the current-sense signal to prevent false-triggering caused by switching noise. The moderate switching frequency (165 kHz) allows good efficiency with moderate-cost and moderate-performance parts.

The transformer (T1) is wound on a Coiltronics SG4 gapped core with AL = 75 nH/T². The primary winding is eight turns of #22AWG, so the primary inductance is 4.8 µH. The secondary winding is 64 turns of #28AWG, so the turns ratio is 1:8. The switch duty cycle is approximately 55%, which gives close to the optimal power transfer for a given magnetic volume.

A lower cost, fast-recovery diode (D1) is used instead of a Schottky diode because the lower switching frequency minimizes the impact of the reverse-recovery time on the switching efficiency. Also, the high-output voltage minimizes the advantage of a lower forward-voltage diode. Because the output voltage is negative, the feedback must be inverted by an op amp (U2) to match the switching controller (U1). D2 protects the inverting input from being pulled negative. R3 and C5 provide the dominant pole for feedback-loop stability.

-24- and -100-V Output from +5.5-V Supply

The circuit in Figure 2 generates -24 V at 150 mA and -100 V at 50 mA for CPE from a +5.5-V minimum input provided by a wall adapter. The MAX668 (U1) has an internal linear regulator that generates a +5-V rail for the gate-drive voltage. The MOSFET (Q1) is selected to switch 7-A peak current with VGS = 4.5 V. Snubbing the flyback voltage is not necessary because the leakage inductance is low and the breakdown voltage of Q1 is more than two times the reflected output voltage. R5 in series with the gate of Q1 slows down the turn-on time to minimize the switching noise seen by the current-sense amp.

The transformer (T1) is an off-the-shelf unit from Coiltronics that allows fast circuit development. A custom transformer can be designed and optimized for volume production. The primary inductance is 3.8 µH, and the turns ratio is 1:1:3. This makes the duty cycle approximately 80% instead of 50% for optimal power transfer. The result is higher peak current compared with an optimized transformer.

The -24-V output is fed back through the op amp inverter (U2), regulating the output to ±1% directly. The -100-V output uses the transformer-turns ratio for regulation. This works as long as the power output from the -100-V supply is not significantly more than the power output from the fed-back voltage. The typical application does not require tight regulation of the -100-V on-hook voltage, and so ±10% is sufficient.

-24- and -48-V Output from +12-V Supply

The circuit of Figure 3 generates -24 V at 50 mA and -48 V at 100 mA from a +12-V nominal input. It demonstrates the optimal turns ratio for power transfer in the transformer. The primary to -24-V secondary turn ratio is 1:2, and the primary to -48-V secondary turns ratio is 1:4. Therefore, the switching regulator operates at 50% duty cycle. The -48-V output is fed back to the controller for regulation. The -24-V output is regulated to ±5% by the turns ratio and close-coupling of the secondary windings.

-24- and -72-V Output with Split Feedback from +12-V Supply

The circuit in Figure 4 generates -24 V at 400 mA and -72 V at 100 mA from a +12-V nominal input. Both outputs are regulated to ±5% under all combinations of line and load by splitting the feedback between the two outputs. The trade-off is to give up a little tolerance on the off-hook voltage for tighter tolerance on the on-hook voltage. The feedback ratios are based on the relative output powers. The -24-V output delivers 4/7th of the maximum output power, so the feedback resistor is scaled to supply 4/7th of the current required for regulation. Similarly, the -72-V output delivers 3/7th of the maximum output power, so the feedback resistor supplies 3/7th of the current required for regulaton.

The transformer is a custom design for this application. The leakage inductance, which represents the imperfect coupling between the primary and secondary windings, is large enough to require snubbing of the primary flyback voltage to prevent breakdown of the switching transistor. R8 and C8 slow down the switch transition and dissipate some of the energy in the leakage inductance to limit the maximum flyback voltage. The Coiltronics Versa-Pac line of transformers are tri-filar wound for maximum coupling (i.e., three wires are wound in parallel), so leakage inductance is minimized. The trade-off is decreased flexibility in the turns ratio and lower isolation-voltage rating between the primary and secondary windings.

Isolated +3.3-, -24-, and -72-V Output from +5-V Supply

The circuit of Figure 5 generates isolated +3.3 V at 100 mA, -24 V at 100 mA, and -72 V at 25 mA from a nominal +5-V input. Isolation is required when the input voltage is not isolated from the line (either electric utility or telephone system power). The transformer isolates the +5-V input from the output voltages. The +3.3-V output is generated from an extra secondary winding. A linear post-regulator is required because of the wide voltage ratio between +3.3 and -24 V used for feedback. The -72-V output regulation is not critical and relies on the turns ratio and close-coupling to the -24-V secondary winding.

The isolated feedback from the -24-V output is implemented with a shunt regulator and an optoisolator. The shunt regulator combines a voltage reference and an error amplifier to generate a current-error signal. The error current drives the photodiode in the optoisolator, which modulates and isolates the current in the phototransistor.

The optoisolator is selected for nominally 100% current-transfer-ratio at 10 mA. The error current is converted to an error voltage through R4. R4 and C13 create a pole in the loop response that limits the loop bandwidth to 2.8 kHz. The loop compensation must take into account the signal delay of the optoisolator as well as the additional gain of the shunt regulator combined with the error amp in the MAX668.