Figure 1 depicts a conceptual schematic of what is generally accepted to be the standard two-stage amplifier topology. Q1 and Q2, together with the current source I1, form a differential pair which is loaded by a Widlar current mirror (Q3 and Q4). These parts make up the first stage; as noted above it is a transconductance gain stage.
Figure 1: Conceptual schematic of the standard two-stage amplifier topology.
The second gain stage with transimpedance behaviour is formed by Q5, Q6 and I2. C1 is the Miller compensation capacitor; it defines, together with the transconductance of the first stage, amplifier open-loop gain at high frequencies and hence stability margins [1,2,3]. Towards high frequencies the compensation capacitor transfers global loop gain to local second stage feedback.
This feedback reference voltage connection of the Miller compensation loop creates several problems. One of these is a reduction in power supply rejection; as the base of Q5 is referenced to the negative supply by the base-emitter junctions of Q5 and Q6, the ripple of the negative power supply rail voltage is superposed to the intended output signal. This effect is reduced by the global feedback loop. The rail injection hence increases towards high frequencies,where less global feedback is available to suppress the effect.
In other words, available amplifier loop gain sets an upper limit to power supply rejection; with standard Miller compensation power supply rejection decreases at 20 dB per decade, and becomes 0 dB at the unity loop gain frequency. With a unity loop gain frequency of 700 kHz, which can be considered a typical figure for audio power amplifiers, power supply rejection is hence limited to about 30 dB at 20 kHz.2 This estimate ignores second-order effects which depend on the exact implementation, but the result is still useful and valid for a rough analysis. Note that for the positive power supply there is no such primary injection path; the power supply rejection will be largely independent of frequency, and thus is usually much less critical.
Power supplies of power amplifiers are typically unregulated for efficiency reasons [1,2]. This means that they carry substantial ripple related both to the mains frequency and harmonics of the output signal; it is highly undesirable that this ripple content be superposed to the audio signal.
Yet it is difficult to give exact figures for the needed power supply rejection, as this depends on the expected performance level, output power and power supply design details. However it is unlikely that the above quoted figure of 30 dB at 20 kHz is sufficient for any quality power amplifier. Also note that ripple related to the harmonics of the output signal extends well above the audio frequency range; hence the region of high power supply rejection should reach at least up to 100 kHz.
To improve the power supply rejection of the basic two-stage topology from figure 1, several solutions have been presented in the literature. The simplest one uses a RC low pass filter, inserted between power output stage and transimpedance stage, to reduce the ripple content on the power supply before it feeds the transimpedance stage.
To be appreciably effective, the low pass filter requires a rather large time constant. If the capacitor shall not become unreasonably large, the resistor must have a sufficiently high value. The inevitable DC voltage drop across this resistor will reduce maximum output voltage swing in many cases. This is further exacerbated by the fact that Q6 will need, in a practical implementation, an emitter resistor for current limiting. This further reduces available output voltage swing.
To avoid this issue, a separate additional negative power supply for the small-signal stages has been used [1,2]. This supply is, for example, derived from an additional mains transformer secondary winding and has rather low current requirements; hence it can easily be arranged to have both sufficient voltage and very low ripple.While the cost of such a solution is modest (at least in the context of a commercial design where a custom power transformer is usually specified anyway), it remains a rather inelegant brute force solution.
Conceptually more pleasing is the use of Ahuja compensation [1,4]. By means of a cascode the input node of the transimpedance stage is referenced to ground rather than a power supply rail. This removes the basic power supply injection route, however in many cases second-order effects make this arrangement less effective than the previously discussed means. Furthermore certain implementations can suffer from local instability of the Miller compensation loop (which now includes at least one additional transistor), or may contribute significantly to the voltage noise of the amplifier.
2 Note that this figure is related to the output, not the input, of the amplifier. This is unlike the figures quoted in operational amplifier data sheets. Figures related to the output show a value which is lower by the noise gain of the used amplifier configuration - typically 20–30 dB for audio power amplifiers.